Present code division multiple access (CDMA) systems are characterized by simultaneous transmission of different data signals over a common channel by assigning each signal a unique code. This unique code is matched with a code of a selected receiver to determine the proper recipient of a data signal. These different data signals arrive at the receiver via multiple paths due to ground clutter and unpredictable signal reflection. Additive effects of these multiple data signals at the receiver may result in significant fading or variation in received signal strength. In general, this fading due to multiple data paths may be diminished by spreading the transmitted energy over a wide bandwidth. This wide bandwidth results in greatly reduced fading compared to narrow band transmission modes such as frequency division multiple access (FDMA) or time division multiple access (TDMA).
Previous studies have shown that multiple transmit antennas may improve reception by increasing transmit diversity for narrow band communication systems. In their paper New Detection Schemes for Transmit Diversity with no Channel Estimation, Tarokh et al. describe such a transmit diversity scheme for a TDMA system. The same concept is described in A Simple Transmitter Diversity Technique for Wireless Communications by Alamouti. Tarokh et al. and Alamouti, however, fail to teach such a transmit diversity scheme for a WCDMA communication system.
New standards are continually emerging for transmit diversity of next generation wideband code division multiple access (WCDMA) communication systems as described in Provisional U.S. Patent Application No. 60/116,268, filed Jan. 19, 1999, and incorporated herein by reference. These WCDMA systems are coherent communications systems with pilot symbol assisted channel estimation schemes. These pilot symbols are transmitted as quadrature phase shift keyed (QPSK) known data in predetermined time frames to any receivers within range. The frames may propagate in a discontinuous transmission (DTX) mode. For voice traffic, transmission of user data occurs when the user speaks, but no data symbol transmission occurs when the user is silent. Similarly for packet data, the user data may be transmitted only when packets are ready to be sent. The frames are subdivided into sixteen equal time slots of 0.625 milliseconds each. Each time slot is further subdivided into equal symbol times. At a data rate of 32 KSPS, for example, each time slot includes twenty symbol times. Each frame includes pilot symbols as well as other control symbols such as transmit power control (TPC) symbols and rate information (RI) symbols. These control symbols include multiple bits otherwise known as chips to distinguish them from data bits. The chip transmission time (TC), therefore, is equal to the symbol time rate (T) divided by the number of chips in the symbol (N).
A mobile unit must initially receive and synchronize with data frames transmitted by one or more remote base stations. Each base station continually transmits broadcast channel (BCH) data over the primary common control physical channel (PCCPCH) to identify itself to mobile units within the cell. Referring to FIG. 1, there is a simplified block diagram of a typical diversity transmitter of the prior art. The transmitter simultaneously transmits primary and secondary synchronization codes on respective primary (P-SCH) 150 and secondary (S-SCH) 160 channels to uniquely identify each base station signal received by the mobile unit. Circuits 156 and 166 modulate the gain of these synchronization codes in response to respective gain factors GP-SCH on lead 154 and GP-SCH on lead 164. Circuit 170 adds the synchronization codes and applies them to time switch (TSW) 174 via lead 172. Time switch 174 selectively applies the synchronization codes to switches SW0 134 and SW1 136 in response to the control signal at lead 140 as indicated by inset 190. These P-SCH and S-SCH codes are transmitted as symbol 300 (FIG. 3) in time slot 1.
Broadcast channel data (BCH) for the PCCPCH are applied to channel encoder 108 via lead 106 (FIG. 1). Interleaver circuit 110 applies the BCH data to space-time transmit diversity (STTD) encoder circuit 112. The STTD encoder produces encoded output data at lead 114 for the transmit antenna (Ant 1) and at lead 116 for the diversity antenna (Ant 2). Multiplex circuit 118 produces this STTD encoded BCH data on leads 120 and 122 at a time corresponding to data symbols 302 of time slot 1 (FIG. 3). The BCH data are modulated by spreading and scrambling codes on lead 124 and applied to switches SW0 134 and SW1 136. These switches SW0 and SW1 selectively multiplex SCH data with BCH data and pilot symbols in response to a control signal on lead 138 as shown at inset 190. The BCH data at lead 180 are then applied to the transmit antenna (Ant 1), and the data at lead 182 is applied to the diversity antenna (Ant 2).
Pilot symbol data for the PCCPCH are applied to lead 100. Diversity circuit 102 generates an open loop transmit diversity (OTD) symbol pattern at lead 104 for the diversity antenna. This OTD pattern together with the pilot symbol pattern for the transmit antenna is shown at TABLE I for each of the sixteen time slots in a frame. By way of comparison, the STTD pilot symbol pattern for diversity antenna (Ant 2) transmission on the dedicated physical data channel (DPDCH) is also shown. The pilot symbols at leads 100 and 102 are applied to multiplex circuit 118. Multiplex circuit 118 selectively applies the pilot symbols at leads 100 and 102 to leads 120 and 122, respectively, at a time corresponding to pilot symbols 304 of time slot 1 (FIG. 3). Thus, multiplex circuit 118 multiplexes STTD encoded data symbols 302 with OTD encoded pilot symbols 304. The pilot symbols at leads 120 and 122 are then modulated with spreading and scrambling code. These modulated pilot symbols at leads 130 and 132 are further multiplexed with SCH data by switches 134 and 136, respectively, in response to the control signal at lead 138 as shown at inset 190. The resulting pilot symbols are applied to transmit and diversity antennas via leads 180 and 182, respectively.
TABLE 1TRANSMIT ANTENNASTTD ANT 2OTD ANT 2SLOTB1S1B2S2B1−S2*−B2S1*B1S1−B2−S211111111111010010111100002111111011111001011110010311011101111100001101001041110110111110011111000105111011111101001111100000611101111110100111110000071101110011100000110100118111011011111001111100010911111100111000101111001110110111011111000011010010111111111011000010111100011211011101111100001101001013110011011111000111000010141110110011100011111000111511011100111000001101001116110011001110000111000011
Turning now to FIG. 2, there is a block diagram showing signal flow in an OTD encoder 102 of the prior art for pilot symbol encoding of the transmitter of FIG. 1. The pilot symbols are predetermined control signals that may be used for channel estimation and other functions as will be described in detail. The OTD encoder 102 receives pilot symbols B1, S1, B2 and S2 at symbol times T-4T, respectively, on lead 100. These pilot symbols are applied to the transmit antenna (Ant 1)via multiplex circuit 118 and switch SW0 134 as previously described. The OTD encoder 102 simultaneously produces pilot symbols B1, S1, −B2 and −S2 at symbol times T-4T, respectively, at lead 104 for the OTD diversity antenna (Ant 2). The pilot symbol pattern for the transmit and OTD diversity antennas is shown at TABLE I for the sixteen time slots of a frame. Each symbol includes two bits representing a real and imaginary component. An asterisk indicates a complex conjugate operation or sign change of the imaginary part of the symbol. Pilot symbol values for the first time slot for the transmit antenna at lead 104, therefore, are 11, 11, 11 and 11. Corresponding pilot symbols for the second antenna at lead 104 are 11, 11, 00 and 00.
The bit signals rj(i+τj) of these symbols are transmitted serially along respective paths 208 and 210. Each bit signal of a respective symbol is subsequently received at a remote mobile antenna 212 after a transmit time τ corresponding to the jth path. The signals propagate to a despreader circuit (FIG. 6) where they are summed over each respective symbol time to produce input signals Rj1, Rj2, Rj3 and Rj4 corresponding to the four pilot symbol time slots and the jth of L multiple signal paths.
The input signals corresponding to the pilot symbols for each time slot are given in equations [1-4]. Noise terms are omitted for simplicity. Received signals Rj1, Rj2, Rj3and Rj4 are produced by respective pilot symbols B1, S1, B2 and S2. Average channel estimates αj^1, and αj^2 over the four pilot symbols for each antenna are obtained from a product of each received signal and a complex conjugate of its respective pilot symbol as in equations [5] and [6].Rj1=(αj1+αj2)B1   [1]Rj2=(αj1+αj2)S1   [2]Rj3=(αj1−αj2)B2   [3]Rj4=(αj1−αj2)S2   [4]αj^1=(B1*Rj1+S1*Rj2+B2*Rj3+S2*Rj4)/4   [5]αj^2=(B1*Rj1+S1*Rj2−B2*Rj3−S2*Rj4)/4   [6]
Referring now to FIG. 4, there is a simplified diagram of a mobile communication system of the prior art. The mobile communication system includes an antenna 400 for transmitting and receiving external signals. The diplexer 402 controls the transmit and receive function of the antenna. Multiple fingers of rake combiner circuit 404 combine received signals from multiple paths. Symbols from the rake combiner circuit 404, including pilot symbol signals, are applied to a bit error rate (BER) circuit 410 and to a Viterbi decoder 406. Decoded symbols from the Viterbi decoder are applied to a frame error rate (FER) circuit 408. Averaging circuit 412 produces one of a FER and BER. This selected error rate is compared to a corresponding target error rate from reference circuit 414 by comparator circuit 416. The compared result is applied to bias circuit 420 via circuit 418 for generating a signal-to-interference ratio (SIR) reference signal on lead 424.
Pilot symbols from the rake combiner 404 are applied to the SIR measurement circuit 432. The SIR measurement circuit produces a received signal strength indicator (RSSI) estimate from an average of received pilot symbols. The SIR measurement circuit also produces an interference signal strength indicator (ISSI) estimate from an average of interference signals from base stations and other mobile systems over many time slots. The SIR measurement circuit produces an SIR estimate from a ratio of the RSSI signal to the ISSI signal. This SIR estimate is compared with a target SIR by circuit 426. This comparison result is applied to TPC command circuit 430 via circuit 428. The TPC command circuit 430 sets a TPC symbol control signal that is transmitted to a remote base station. This TPC symbol instructs the base station to either increase or decrease transmit power by 1 dB for subsequent transmission.
Turning now to FIG. 5, there is a diagram showing a weighted multi-slot averaging (WMSA) circuit 732 of the prior art for channel estimation. In operation, a signal buffer circuit 706 (FIG. 7) receives individual frames of data having a predetermined time period of 10 milliseconds. Each frame of the PCCPCH is subdivided into sixteen equal time slots of 0.625 milliseconds each. Each time slot, for example time slot 528, includes a respective set of pilot symbols 520 and data symbols 529. The WMSA circuit (FIG. 5) samples pilot symbols from preferably 6 time slots for a Doppler frequency of less than 80 Hz and from preferably 4 time slots for a Doppler frequency of 80 Hz or more. These sampled pilot symbols are multiplied by respective weighting coefficients α1, through αN and combined by circuit 526 to produce a channel estimate. This channel estimate is used to correct the phase of received data symbols in time slot 527 estimate for a respective transmit antenna.
Referring now to FIG. 6, there is a despreader circuit of the prior art. Received signals from mobile antenna 212 propagate to the despreader circuit where they are summed over each respective symbol time to produce output signals Rj1 and Rj2 corresponding to the jth of L multiple signal paths as previously described. The despreader circuit receives the ith of N chip signals per symbol together with noise along the jth of L multiple signal paths at a time τj after transmission. Both here and in the following text, noise terms are omitted for simplicity. This received signal rj(i+τj) at lead 600 is multiplied by a channel orthogonal code signal Cm(i+τj) at lead 604 that is unique to the receiver. Each chip signal is summed over a respective symbol time by circuit 608 and produced as first and second output signals Rj1 and Rj2 on leads 612 and 614 as in equations [1-2], respectively. Delay circuit 610 provides a one-symbol delay T so that the output signals are produced simultaneously.
This arrangement advantageously provides additional gain at the mobile communication system by multiple path transmit antenna diversity from a remote base station. The mobile unit, however, must be compatible with base stations having a single transmit antenna as well as base stations having transmit antenna diversity. A problem arises, therefore, when the mobile communication system is initially powered up or when it passes from one cell to another cell. The mobile unit must not only determine which of several base signals offers a preferable signal strength. It must also determine whether the base station offers transmit antenna diversity. If the mobile unit incorrectly decodes a received signal and assumes no transmit diversity, it loses the improved gain of transmit diversity. Alternatively, if the mobile unit incorrectly decodes a received signal and assumes transmit diversity, multiple fingers of the rake combiner circuit 404 contribute noise to the received signal.